Misalignment-Tolerant Series Hybrid with Active Adjustable Constant Current and Constant Voltage Output Wireless Charging System

Abstract

This paper presents a series hybrid wireless charging system with an active adjustable circuitry offering constant current and constant voltage output characteristics. The series hybrid system consists of the inductor–capacitor–capacitor (LCC) and series-series (SS) networks are used for improving charging pad misalignment tolerance. An active switch is employed to provide an adjustable CC and CV output for different battery charging stages. To demonstrate the performance of the proposed method, a 310 W prototype was built. A systematic optimization in the parameter of the proposed topology to achieve relative constant output was analyzed within a certain range of the designed operating region. The experimental results indicate that the output current fluctuation is less than 5% with load variations, and the output voltage fluctuation is less than 5% with load varying from 19 to 70 Ω, as the pick-up pads misaligned within 50% of the pad outer diameter.

1. Introduction

Wireless power transfer (WPT) systems using an alternating magnetic field to transfer power across a relatively large air gap have been adopted in numerous industrial and commercial applications. In comparison to traditional plug-in charging technology, WPT is safe offering galvanic isolation, maintenance free with dirt and moisture environments [1,2,3,4,5,6], and has great potential to be used in, for example, electric vehicle (EV) charging, portable electronics, autonomous underwater vehicles (AUVs), and implantable biomedical devices [7,8,9,10,11], etc.

As for practical battery charging applications, the equivalent resistance of the battery varies significantly when considering the entire charging process. Thus, charging with a wide range of loading conditions to achieve the constant current (CC) output and constant voltage (CV) output is essential for most of the li-ion battery charging applications [12]. Specific to wireless charging scenarios, misalignment between magnetic couplers results in variation of the self and mutual inductance, and poses a negative impact on the current, voltage outputs, and transfer efficiency. Thus, the motivation of this paper is to design a WPT system, which is able to provide CC output and CV output with a wide range of loads and high misalignment tolerance between the primary and pickup magnetic couplers.

In the past few years, many approaches have been investigated to realize the desired current or voltage under different loading conditions for a WPT system. The DC-DC converter at the primary side or the secondary side is employed to regulate the current or voltage output [13,14]. However, this additional cascading DC-DC converter will result in extra converter loss and cost, and need extra space for setting up the device. Variable frequency control and phase shift control of the WPT system can also be used to realize the CV or CC output against load and coupling variations [15,16,17]. However, variable frequency control may result in the frequency bifurcation phenomenon in a certain loading condition with a reduced power transmission capability [18]. In addition, the phase shift control has difficulty realizing the input zero phase angle (ZPA). Furthermore, these closed-loop control methods all require feedback signals from the pick-up side, which may result in control failure when wireless communication is disturbed. To solve these drawbacks, compensation topologies to obtain relative CC or CV output without the complicated control approaches have also been investigated. A variable coil structure for WPT system CC and CV output, which consists of two DD coils and one Q coil on the transmitter side, is systematically analyzed in [19]. In [20], a variable-parameter T-circuit on the primary side is used to regulate the output current and voltage, reducing the number of passive components. However, these variable structures on the transmitter side allow the WPT system to operate with the desired CC and CV output at the expense of extra wireless communication, which can result in additional system cost and control failure due to external interference. In order to solve the deficiency of wireless communication, some new variable compensation topologies at the receiver side, such as series–series (SS) and series–inductor–capacitor–capacitor (S-LCC), or inductor–capacitor–capacitor–series (LCC-S) and LCC-LCC topologies for CC and CV outputs, are introduced in [21,22]. Nonetheless, all the compensation topologies aforementioned can achieve desired CC and CV outputs when the magnetic couplers are perfectly aligned at a fixed position. Otherwise, the misalignment may seriously affect the current and voltage output due to parameter variations.

The misalignment of the coupling pads is expected in practical applications, which may result in decreasing power transmission capability, reducing the transmission efficiency, and instability of the system. Thus, to maintain constant power when the misalignment occurs, a practical WPT application is necessary. Numerous approaches, such as control techniques, proper magnetic structures, and hybrid circuit topologies, have been used to mitigate this problem.

Many new control strategies have been proposed [23,24,25,26] to maintain stable outputs under pad misalignment conditions. However, these controllers usually need wireless communication to obtain the feedback signals from the secondary side, and regulate the power flow. The communications may result in instability of the WPT systems, regardless of the extra volume and cost. In order to solve these defects, considerable efforts have been made to design proper magnetic couplers, such as three-dimensional (3D) quadrature-shaped coils [27], bipolar and double-D (DD) coils [28], quadruple-D quadrature pads (QDQP) [29], and an unsymmetrical coupling structure [30]. These magnetic couplers can provide a relatively uniform magnetic field distribution to improve the misalignment tolerance. Meanwhile, hybrid topologies, which are an alternative method used to improve the misalignment tolerance, have recently been introduced in [29,31,32,33]. For example, in [29], an input-parallel–output-series hybrid system was built, which consists of LCC-S and S-LCC topologies. The new compensation topology can maintain the output voltage within 5% fluctuations, when 50% longitudinal misalignment occurs. The series-hybrid and parallel-hybrid topologies are presented in [31,32], with the combination of LCC-LCC and SS topologies, which were connected in series and parallel types of the primary and pick-up sides, separately, to achieve constant power throughput. A family of hybrid WPT topologies is discussed in detail in [33], which can extend the tolerant misalignment. However, all these approaches above cannot achieve both CC output and CV output with high misalignment.

An input-parallel–output-series hybrid and configurable system is proposed in [34], which can provide relatively constant CC and CV outputs, within 50% longitudinal misalignment. However, when the pick-up pads move out of the operating region, the current of the parallel SS compensation network in the primary side may increase too much to break down the inverter, which is caused by the decrease in reflected impedance due to the decreasing coupling. Although using a closed-loop controller can suppress this increase current and protect the inverter, it may significantly reduce the system efficiency and reliability.

This paper presents a novel series hybrid and configurable compensation topology to achieve CC and CV outputs with high misalignment tolerance. Additionally, the main contributions of this paper are summarized as follows:

(1) The proposed series hybrid compensation topology can achieve load-independent CC output, and realize operation without pickup side coils, which can overcome the aforementioned drawbacks associated with [34], and limit the current of the inverter without extra controllers. In addition, a set of parameter optimization is proposed to improve the constancy in output power when misalignment occurs.

(2) A variable structure T-circuit is employed to transfer CC output to CV output without an additional cascade converter, and owns a native load-independent character, which can simplify the control schemes.

Specifically, Section 2 analyzes the mathematical model of the series hybrid and configurable topology. The changes in the inductances of the DDQ pads between primary and pick-up sides, and the parameter optimization are presented in Section 3. In Section 4, a 310 W prototype that was built to verify the theoretical analysis is described. Finally, the conclusion is drawn in Section 5.

2. Theoretical Analysis

Figure 1 shows the proposed series hybrid and configurable topology. The high frequency inverter and full-bridge rectifier consists of four MOSFETs (Q₁–Q₄) and four diodes (D₁–D₄), respectively. An inductor L₀ and capacitors C₀ and C₁ (L₅, C₅ and C₂) constitute the LCC-LCC compensation topology, while capacitor C₃ (C₄) is part of the SS compensation topology. LCC-LCC and SS topologies of the primary and pick-up sides are both connected in series, together making up the LCC-S series hybrid topology, and the configurable topology is formed with inductors L_f1 and L_f2 and capacitor C_f, together with switches S₁ and S₂. M₁₂ and M₃₄ are the main mutual inductances, while M₁₃, M₁₄, M₂₃, and M₂₄ are the cross mutual inductances. The analysis of the proposed system is based on the fundamental harmonic approximation method and the influence of the higher harmonics is ignored.

2.1. Analysis of the Series Hybrid Topology

Figure 2 illustrates the series hybrid topology, which is driven by a resonant inverter with angular frequency ω. In order to tune the LCC-S series hybrid compensation network in the primary and pick-up sides to the constant resonant frequency, the resonant parameters are designed to satisfy the following equations:

$$\{\begin{array}{l}{\omega }^2{L}_0{C}_0={\omega }^2{L}_1\frac{C_0{C}_1}{C_0+C_1}=1\\ {\omega }^2{L}_5{C}_5={\omega }^2{L}_2\frac{C_2{C}_5}{C_2+C_5}=1.\\ {\omega }^2{L}_3{C}_3={\omega }^2{L}_4{C}_4=1\end{array}$$

(1)

According to Figure 2, Kirchhoff’s voltage law is used to describe the relationship of voltages in the series hybrid circuit; therefore, we can obtain:

$$\left[\begin{array}{cccc}{Z}_{00}& Z_{01}& Z_{02}& Z_{03}\\ Z_{10}& Z_{11}& Z_{12}& Z_{13}\\ Z_{20}& Z_{21}& Z_{22}& Z_{23}\\ Z_{30}& Z_{31}& Z_{32}& Z_{33}\end{array}\right]\left[\begin{array}{c}{\dot{I}}_0\\ {\dot{I}}_1\\ {\dot{I}}_2\\ {\dot{I}}_3\end{array}\right]=\left[\begin{array}{c}{\dot{U}}_{in}\\ 0\\ 0\\ 0\end{array}\right]$$

(2)

where:

$Z_{00}=j\omega L_0+{\left(j\omega C_0\right)}^{-1}+j\omega L_3+{\left(j\omega C_3\right)}^{-1}$, $Z_{01}=-{\left(j\omega C_0\right)}^{-1}+j\omega M_{13}$, $Z_{02}=-j\omega M_{23}$, $Z_{03}=j\omega M_{34}$, $Z_{10}=j\omega M_{13}-{\left(j\omega C_0\right)}^{-1}$, $Z_{11}=j\omega L_1+{\left(j\omega C_0\right)}^{-1}+{\left(j\omega C_1\right)}^{-1}$, $Z_{12}=-j\omega M_{12}$, $Z_{13}=j\omega M_{14}$, $Z_{20}=j\omega M_{23}$, $Z_{21}=j\omega M_{12}$, $Z_{22}=j\omega L_2+{\left(j\omega C_2\right)}^{-1}+{\left(j\omega C_5\right)}^{-1}$, $Z_{23}={\left(j\omega C_5\right)}^{-1}+j\omega M_{24}$, $Z_{30}=j\omega M_{34}$, $Z_{31}=j\omega M_{14}$, $Z_{32}=-{\left(j\omega C_5\right)}^{-1}-j\omega M_{24}$, $Z_{33}=j\omega L_5+{\left(j\omega C_5\right)}^{-1}+j\omega L_4+{\left(j\omega C_4\right)}^{-1}+R_{AB}$.

In order to simplify the analysis, only the main couplings (M₁₂ and M₃₄) are taken into consideration, while the cross couplings (M₁₃, M₁₄, M₂₃, and M₂₄) can be ignored by designing proper coupling structures, which will be discussed in Section 3. Hence, the current of the inverter, the current of the L₁ transmitter, the current of the L₂ receiver, and the current of the R_AB load are obtained as:

$$\{\begin{array}{l}{\dot{I}}_0=\frac{{\dot{U}}_{in}}{{\omega }^2}\frac{M_{12}^2{R}_{AB}}{{(L_0{L}_5+M_{12}{M}_{34})}^2}\\ {\dot{I}}_1=\frac{{\dot{U}}_{in}}{j\omega }\frac{L_5}{L_0{L}_5+M_{12}{M}_{34}}\\ {\dot{I}}_2=\frac{{\dot{U}}_{in}}{{\omega }^2}\frac{L_0{M}_{12}^{}{R}_{AB}}{{(L_0{L}_5+M_{12}{M}_{34})}^2}\\ {\dot{I}}_3=\frac{{\dot{U}}_{in}}{j\omega }\frac{M_{12}}{L_0{L}_5+M_{12}{M}_{34}}\end{array}$$

(3)

Then, according to the current ${\dot{I}}_0$ of the inverter, the input equivalent impedance Z_in of the system can be deduced as:

$$Z_{in}=\frac{{\omega }^2{\left(L_0{L}_3+M_{12}{M}_{34}\right)}^2}{M_{12}{}^2{\mathrm{R}}_{AB}}$$

(4)

According to (3) and (4), the current ${\dot{I}}_0$ is in the same phase with the inverter output voltage ${\dot{U}}_{in}$, and the input impedance of the system is pure resistance. When the whole system works in the full tuned condition, the input reactive power is zero, which can improve the transmission efficiency of the system. In addition, the current ${\dot{I}}_0$ and the input equivalent impedance Z_in clearly indicate that the inverter current is close to zero when the pick-up sides move far away, which means the mutual inductance is close to zero. Therefore, the series hybrid system can avoid the extreme situation that the inverter output current is too high to burn down due to the no-load operation of the parallel SS structure, and improve the reliability of the system.

Equation (3) shows that the output current ${\dot{I}}_3$ is load independent and lags the inverter output voltage by 90°.

2.2. Analysis of the Configurable Topology

Figure 3 shows the equivalent circuit of the configurable topology is driven by a current source, because the series hybrid topology circuit can obtain load-independent constant current output. When the function switch S₁ is ON and S₂ is OFF, the system provides CC output, and the output current ${\dot{I}}_{CD}$ is shown as:

$${\dot{I}}_{CD}={\dot{I}}_3$$

(5)

The input equivalent impedance Z_cc-AB is expressed as:

$$Z_{cc-AB}=R_{CD}$$

(6)

When the S₁ is OFF and S₂ is ON, the system provides CV output, as is shown in Figure 4. The relationship between inductors L_f1 and L_f2 and capacitor C_f satisfies the following equations, i.e.,

$$\{\begin{array}{l}j\omega L_{f1}+\frac{1}{j\omega C_f}=0\\ j\omega L_{f2}+\frac{1}{j\omega C_f}=0\end{array}$$

(7)

Then, according to Thevenin–Norton theorems, the input voltage shown in Figure 5 can be described by:

$${\dot{U}}_{AB}=\frac{{\dot{I}}_3}{j\omega C_f}$$

(8)

Equation (6) indicates that the inductor L_f2 and capacitor C_f form the resonant network tank; therefore, the output voltage is:

$${\dot{U}}_{CD}=\frac{{\dot{I}}_3}{j\omega C_f}$$

(9)

The input equivalent impedance Z_cv-AB is expressed as:

$$Z_{cv-AB}=\frac{L_{f1}}{C_f}{R}_{CD}$$

(10)

2.3. The Proposed Combination of Series Hybrid and Configurable Topology

Figure 6 shows the equivalent circuit with the series hybrid and the configurable topology. Comparing Figure 1 with Figure 6, the capacitor C_E is used to substitute the inductor L₀ (L₅) and capacitor C₃ (C₄). The component C_E can be expressed by:

$$j\omega L_0+\frac{1}{j\omega C_3}=\frac{1}{j\omega C_E}$$

(11)

or:

$$j\omega L_5+\frac{1}{j\omega C_4}=\frac{1}{j\omega C_E}$$

(12)

From (3) and (5), when the system operates in CC output, the output current is:

$${\dot{I}}_{CD}=\frac{{\dot{U}}_{in}}{j\omega }\frac{M_{12}}{L_0{L}_5+M_{12}{M}_{34}}$$

(13)

Additionally, the corresponding input impedance Z_in-cc is expressed by:

$$Z_{in-cc}=\frac{{\omega }^2{\left(L_0{L}_5+M_{12}{M}_{34}\right)}^2}{M_{12}{}^2{\mathrm{R}}_{CD}}$$

(14)

From (3) and (8), when the system operates in CV output, the output voltage is:

$${\dot{U}}_{CD}=-\frac{{\dot{U}}_{in}{M}_{12}}{{\omega }^2{C}_f(L_0{L}_5+M_{12}{M}_{34})}$$

(15)

Additionally, the corresponding input impedance Z_in-cv is expressed by:

$$Z_{in-cc}=\frac{{\omega }^2{\left(L_0{L}_5+M_{12}{M}_{34}\right)}^2{C}_f{\mathrm{R}}_{CD}}{M_{12}{}^2{L}_{f1}}$$

(16)

According to (12) and (14), the value of L₀L₅/M₁₂ will increase, while the value of M₃₄ will decrease, when the main mutual inductances M₁₂ and M₃₄ decrease with pad misalignment. Thus, the sum of L₀L₅/M₁₂ and M₃₄ can remain relatively constant within a certain range of misalignment when the parameters are properly designed. Then, a constant current and voltage output with pad misalignment can be achieved.

From (13) and (15), it illustrates that the input impedance of the system is pure resistance when the series hybrid and configurable topology works in CC output and CV output, which means the output voltage and current of the inverter can achieve a zero phase angle (ZPA).

3. Design and Implementation of the Hybrid and Configurable Topology

3.1. Coupler Design

According to (3), constant current output can be achieved only when the cross couplings (M₁₃, M₁₄, M₂₃, and M₂₄) are too small to be negligible. Therefore, only the main mutual inductances M₁₂ and M₃₄ are taken into consideration when the magnetic coupler is in misalignment. Additionally, the DDQ and DD structures can satisfy these desired requirements as discussed in [32,33]. Hence, the DDQ structure is used and shown in Figure 7. The Q pad structure is formed by the transmitter L₁ and receiver L₂, while the DD pad structure is formed by the transmitter L₃ and receiver L₄. Meanwhile, the misalignment occurrence between the primary and pick-up pads is unavoidable in the charging system, including X-axis misalignment, Y-axis misalignment, and Z-axis misalignment. All the mutual inductances of the DDQ coils are measured when the pick-up pads are moved along the X-axis, Y-axis, and Z-axis, separately.

As is shown in Figure 8, the main mutual inductances M₁₂ and M₃₄ decrease apparently with the increase of X-axis and Z-axis misalignments, while the cross coupling mutual inductances are too small to be neglected, because the Q pad and the DD pad are symmetrically placed in the primary and pick-up side, so that the amount of the magnetic flux that flows into the Q (DD) pad equals the flows out of it. However, when Y-axis misalignment occurs, the main mutual inductances M₁₂ and M₃₄ and the cross coupling mutual inductances M₂₃ and M₁₄ vary significantly, so the proposed system cannot achieve CC output and CV output in the Y-axis misalignment.

3.2. Parameter Optimization Design

In accordance with the analysis in Section 2, designing proper compensation parameters is of great significance to achieve the relative constant output current and voltage within a certain range of misalignment. In this article, a parameter optimization design method based on inductance L₀ and L₅ is proposed to ensure relative constant current output of the system.

According to Figure 8a of the X-axis misalignment curves, the relationship between M₁₂ and M₃₄ can be approximately regarded as a linear function:

$$M_{34}=a{M}_{12}+b$$

(17)

where a and b are the coefficients. When the structural parameters of the DDQ coil of the system, such as the material, size, and spacing height of the coil, are changed, resulting in the change trend of main mutual inductances M₁₂ and M₃₄, the parameters a and b need to be recalculated, respectively.

From Figure 8a of the X-axis misalignment curves, the calculated parameters a and b are 0.52 and 2.17 × 10⁻⁶, respectively, and the variation range of the main mutual inductance M₁₂ is [14 uH, 30 uH], then the output current can be expressed as:

$$I_3=\frac{U_{in}{M}_{12}}{\omega (L_0{L}_5+M_{12}(0.52M_{12}+2.17\times {10}^{-6}))}$$

(18)

To simplify the analysis, it is assumed that the parameter values of inductance L₀ and L₅ are equal. Figure 9 shows the output current curves of the system with different inductors L₀. It can be found that when the inductor L₀ decreases, the output current of the system will increase. In addition, when the main mutual inductance M₁₂ decreases, the system output current shows a trend of increasing first and then decreasing. In this article, the maximum output current of the system is designed to be 4 A, and the allowable deviation of the current is 5%. In other words, the area in the red region of Figure 9 can meet the constant current output under the condition of 50% X-axis offset. Therefore, the inductors L₀ and L₅ are selected as 16 uH.

According to the measured self-inductance L₁ and L₂ of Q coils, self-inductance L₃ and L₄ of DD coils, system resonant angular frequency ω, and inductance L₀ and L₅ with optimized parameters, the parameters of components, such as capacitors C₀, C₁, C₂, C₃, C₄, and C₅, can be obtained from (1).

4. Experimental Results and Discussion

Figure 10 shows the proposed series hybrid and configurable wireless charging system. The inverter switching devices Q₁–Q₄ MOSFETs use C2M0080120 and the function switching devices S₁–S₂ MOSFETs use SPW47N60C3. An electronic load is used to verify the performance of the CC and CV output. The system parameters are listed in

Table 1. Configurations of the WPT system.

Parameter	Value	Parameter	Value
f	85 kHz	C1	26.6 nF
L1	150.1 uH	C2	220.1 nF
L2	149.8 uH	C3	22.5 nF
L3	156.1 uH	C4	22.5 nF
L4	156.0 uH	C5	26.5 nF
CE	25.0 nF	E	70 V
C0	220.2 nF

.

4.1. CC Output Performance of the System

Figure 11 plots the measured output current varying with the loads and X-axis misalignment. Within 140 mm X-axis misalignment, the output current of the system is between 3.85 and 4.25 A, meeting the requirement of 5% deviation. Under the same misalignment condition, the output current of 15 Ω is largest, and the output current of 18 Ω slightly decreased. If the load becomes lighter, the current fluctuation will exceed the limitation of 5%. In addition, the system output current is minimum when the receiving coil is offset to 140 mm. Additionally, it is clearly found that the load current increases first and then decreases with the increase of the offset distance, which is beneficial to improve the anti-offset ability of the system.

As is shown in Figure 12, the output voltage U_in and the output current I_in of the inverter are almost in the same phase, which indicates that near zero reactive power is achieved. Additionally, the output voltage U_L and output current I_L, indicate that the system can achieve a relative constant current output within 50% X-axis misalignment.

The function switches consist of two anti-series-connected MOSFETs, which are shown in Figure 13. When the S₁ switch is ON and the S₂ switch is OFF, the system works in CC mode. Otherwise, the system works in CV mode. Figure 14 shows the sudden transient waveforms. There are some oscillations when the charging mode changes from CC to CV output. In CC mode, the output current is around 4 A, and the output voltage is about 72 V in CV mode. In addition, the system only takes about 4 ms to reach the new steady state.

4.2. CV Output Performance of the System

As is shown in Figure 15, the output voltage of the system is between 68.5 and 75.5 V within 140 mm X-axis misalignment, meeting the requirement of 5% deviation, when the loads vary from 19 to 70 Ω. In addition, if the load becomes lighter, the voltage fluctuation will exceed the limitation of 5%. Additionally, the system output voltage climbs to the peak when the receiving coil is offset to 80 mm, which is consistent with the trend of the output current.

Figure 16 shows that there is a small phase angle between the voltage and the current of the inverter, which indicates that ZVS is achieved within 50% X-axis misalignment. Additionally, the output voltage U_L and output current I_L indicate that the system can achieve relative constant voltage output within 50% X-axis misalignment.

Figure 17 shows the experiment waveforms when the pick-up sides moved far away. The output voltage of the system is zero, and the current of the inverter is near zero, which indicates that the proposed system can work without the pick-up sides, and increase the stability of the system without any extra controllers.

Figure 18 shows that the efficiency in CC mode is between 91.0% and 89.2%, while the efficiency in CV mode is between 90.5% and 85.2%. In addition, the efficiency drops with the increase of the X-axis misalignment, because the increasing current flows into the L₁ coil, resulting in increasing corresponding conductance loss.

4.3. Comparison with Other Methods

The performance of the proposed approach was compared with methods using control schemes and compensation topologies, as listed in

Table 2. Comparison with methods using control schemes and compensation topologies.

Proposed in	Ref. [14]	Ref. [16]	Ref. [17]	Ref. [19]	Ref. [20]	Ref. [31]	Ref. [32]	Ref. [33]	Ref. [34]	This Work
Control scheme	Yes	Yes	Yes	No	No	No	No	No	No	No
Number of Component	2 + dc-dc converter	2	2 + semi-active rectifier	4	5	8	6	8	12	11
k	0.2	0.2	0.15	0.22	0.17	0.18–0.30	0.15–0.35	0.08–0.16	0.1–0.2	0.1–0.2
Max. power	3 kW	103 W	66 W	105 W	400 W	3.3 kW	3.3 kW	3.5 kW	1 kW	280 W
CC or CV	CC	CC-CV	CC	CC-CV	CC-CV	CC	CC	CV	CC-CV	CC-CV
Peak efficiency	88%	75%	86%	91%	94%	91%	94%	93%	94%	91%
Frequency (kHz)	85.5	85.5	85.5	500	85	85	85	85	85	85
Misalignment tolerance	No	No	No	No	No	Yes	Yes	Yes	Yes	Yes
Wireless communication	Yes	No	No	Yes	Yes	No	No	No	No	No
Operate without pickup pad	\	\	\	\	\	No	Yes	Yes	No	Yes

. Compared with methods using control schemes, such as [14,16,17], the proposed WPT system can realize CC-CV output without an additional control scheme. Compared with methods using variable compensation topologies, such as [19,20], the proposed WPT system can both realize CC-CV output and misalignment tolerance. Among [31,32,33], the hybrid topologies are employed to improve misalignment tolerance. However, they can only achieve CC or CV output. Compared with [34], the proposed WPT system uses fewer components, and can operate without the pickup side, which improves the reliability of the system. Therefore, the proposed WPT system is superior to the other approaches in terms of CC and CV output, misalignment tolerance, and the ability of operating without a pickup pad.

5. Conclusions

In this article, a series hybrid wireless charging system with an active adjustable circuitry is proposed to obtain CC and CV outputs with high misalignment tolerance. The system is combined with series LCC-S topology in the primary side and pick-up side and T-type configurable topology, which can limit the current of the inverter without an extra controller when the pick-up pad moves out of the operating region. Besides, the function switch is employed to transfer CC output to CV output without complicated control schemes. Moreover, a parameter optimization design method is presented to provide high misalignment tolerance. The experimental results demonstrate that the system can maintain the output current fluctuation to be less than 5% with the load varying from 15 to 18 Ω, and the output voltage fluctuation less than 5% with the load varying from 19 to 70 Ω, when the pick-up pads are within 50% misalignment. The results demonstrate the theoretical analysis, and indicate that the series hybrid and configurable wireless charging system offers a reliable solution to wireless EV charging applications.